Multi-mode/multi-band power amplifiers (PAs) are designed to amplify radio frequency (RF) signals that may operate in second generation (2G) wireless access networks that use Gaussian minimum shift keying (GMSK) and enhanced data GSM evolution (EDGE) modulation protocols. Third generation (3G) wireless access networks that use high speed packet access (HSPA) and enhanced high speed packet access (HSPA+) modulation protocols may also be amplified by multi-mode/multi-band PAs. Further still, fourth generation (4G) wireless access networks that use long term evolution frequency division duplex (LTE FDD) and time division duplex (TDD) may also be amplified by multi-mode/multi-band PAs. Other dedicated transmit chain amplifiers are used to support extra bands that operate with a single mode. These dedicated transmit chain amplifiers are sometimes referred to as “bolt-on” PAs. For example, a bolt-on amplifier could be used to support 4G specific bands that are outside the bands that are compatible with an individual multi-mode/multi-band PA.
FIG. 1 is a block diagram of a related art power management system 10 for a multi-mode/multi-band PA 12 and bolt-on PAs 14, all of which are powered by a single power converter 16. The power converter 16 also includes a serial interface 18 that sends and receives data and receives commands for controlling the power management system 10. A typical serial interface is a mobile industry processor interface (MIPI) for communicating with a radio frequency front end (RFFE).
An output filter 20 is coupled between the power converter 16 and a power node 22 for both the multi-mode/multi-band PA 12 and the bolt-on PAs 14. The output filter 20 is an LC type filter that includes an inductor LGMSK/LINEAR that is coupled between the power converter 16 and the power node 22. Also, a capacitor CGMSK/LINEAR is coupled between the inductor LGMSK/LINEAR and ground GND.
The power management system 10 provides power to the multi-mode/multi-band PA 12 for a saturated amplifier mode used to amplify GMSK modulated signals. When GMSK signals are being amplified by the multi-mode/multi-band PA 12, a slow DC-DC mode of the power converter 16 provides a constant direct current (DC) voltage output. In order to meet GMSK requirements that limit a magnitude of ripple current flowing through the inductor LGMSK/LINEAR, an inductance value for the inductor LGMSK/LINEAR is relatively large, ranging from around 1 μH to 2 μH. Moreover, a capacitance value for the capacitor CGMSK/LINEAR is relative large, being around 2 μF in order to meet GMSK spectrum requirements. An integration of the ripple current flowing through the inductor LGMSK/LINEAR divided by the capacitance value of the capacitor CGMSK/LINEAR provides a value for an undesirable ripple voltage that appears across the power node 22 to ground GND. In order to reduce this undesirable ripple voltage, either a large inductance value for the inductor LGMSK/LINEAR is needed or a larger capacitance value for the capacitor CGMSK/LINEAR is needed. Alternately, a high switching frequency for the power converter 16 can be used.
The power management system 10 must also provide power to the multi-mode/multi-band PA 12 and the bolt-on PAs 14 when they are operated in a linear amplifier mode used to amplify signals for 3G and 4G operation. Since the linear amplifier mode is typically less efficient than the saturated amplifier mode, envelope tracking is employed wherein the power converter 16 outputs a supply voltage VCC that is amplitude modulated in synchronization with the amplitude of a signal that is to be amplified by a PA of either the multi-mode/multi-band PA 12 or the bolt-on PAs 14. In this way, efficient operation of the multi-mode/multi-band PA 12 and the bolt-on PAs 14 is increased as the supply voltage follows the amplitude of the signal to be amplified. A power control signal known as VRAMP received by the power converter 16 carries envelope tracking information that is used by a parallel amplifier 24 to control the amplitude modulation of the supply voltage VCC, which is typically on the order of 3V peak-to-peak (pkpk).
A problem arises when operation in envelope tracking mode in that the parallel amplifier 24 will have to provide the relatively large voltage modulation of 3Vpkpk across the capacitor CGMSK/LINEAR that serves as a large decoupling capacitor during the saturated amplifier mode. The problem is that a relatively large modulation current IBYPASS is demanded from the parallel amplifier 24 in order to satisfy the following relationship that assumes a 10 MHz modulation of the supply voltage VCC.IBYPASS=CGMSK/LINEAR*dVCC/dt=2 μF*3Vpkpk*10 MHz=60A
In actuality, the modulation current IBYPASS is lower than 60A since the amplitude modulation provided by the parallel amplifier 24 is typically only around 1 MHz to 5 MHz. Nevertheless, the modulation current IBYPASS remains unacceptably large.
FIG. 2 is a schematic diagram of a system model 26 that is usable to derive a transfer function for the power management system 10 of FIG. 1. The system model 26 includes a multi-level switcher 28 that is controlled by a source voltage VCCI that is responsive to a power control signal VRAMPS—C that equals the power control signal VRAMP without pre-distortion. The system model 26 also includes a PA model 30 that represents a PA under the control of the power management system 10. A resistance RPA in parallel with a current source IPA models a response of the PA to an output current IOUT(S). A voltage VCCPA is developed across the PA in response to the output current IOUT(S).
An output voltage VCC0 represents an output voltage VCC versus the power control signal VRAMP when a switching current ISW(S) output from the multi-level switcher 28 is equal to zero. A transfer function HEQ(S)=VCC0/VRAMPS—C when the switching current ISW(S) output from the multi-level switcher 28 is equal to zero. Extra decoupling capacitors located within the PA are represented in the PA model 30 as a decoupling capacitor CPA. Extra parasitic inductance that results from layout routing of a power line from a VCC node to the PA is represented by a parasitic inductor LPA.
During operation in the envelope tracking mode for LTE, an inductor value for the inductor LGMSK/LINEAR (FIG. 1), and represented by L should be lowered to around 0.5 μH to allow the multi-level switcher 28 to quickly provide current through the inductor LGMSK/LINEAR. In this way, a slew rate limitation is avoided. For example, a current ISW(S) through the inductor LGMSK/LINEAR, and represented by L has a current rate of change dlSW/dt that is given by the following equation.dlSW/dt=(VSW−VCC)/(LGMSK/LINEAR)
As a result, the current rate of change dlSW/dt is limited by an amount of available headroom VSW−VCC, which is usually increased by a value of inductance for the inductor LGMSK/LINEAR. A lower inductance value for LGMSK/LINEAR will result in a higher current rate of change dlSW/dt. Thus, care must be taken not to lower the inductance value for LGMSK/LINEAR such that an instantaneous switching frequency of the switcher is required to be increased.
A more detailed analysis of the power management system 10 can be achieved by considering the various impedances that are represented in the system model 26. For example, an impedance Z1 is made up of the model inductor L that represents the impedance of the inductor LGMSK/LINEAR. An impedance Z2 represents optional filtering components and parasitic inductance between an output of the parallel amplifier 24 (FIG. 1) and a PA represented by the PA model 30. An impedance Z3 is primarily made up of the parasitic inductance LPA in parallel with the decoupling capacitor CPA coupled the resistance RPA. An impedance ZOUT represents the impedance of the parallel amplifier 24 in parallel with a decoupling capacitor located on the VCC node.
The following equations are derived from the system model 26 shown in FIG. 2.VCC/VCC0=(Z1/(Z1+ZOUT))*(Z2+Z3)/(Z2+Z3+ZOUT*Z1/(ZOUT+Z1))VCCPA/VCC0=(Z1/(Z1+ZOUT))*(Z3/(Z2+Z3+ZOUT*Z1/(ZOUT+Z1))VCC/VSW=(ZOUT/(Z1+ZOUT))*(Z2+Z3)/(Z2+Z3+ZOUT*Z1/(ZOUT+Z1))VCCPA/VSW=(ZOUT/(Z1+ZOUT))*(Z3/(Z2+Z3+ZOUT*Z1/(ZOUT+Z1))
A computer simulation of the system model 26 starts with VRAMPS—C set to zero, which forces VCC0 equal to zero. In this way, the slow DCDC mode for saturated amplifier operation such as needed with GSMK is simulated. A typical inductance value for the impedance Z1 is set to about 1 μH, and a decoupling capacitance value for ZOUT is set to around 2.2 μF for simulating the slow DCDC mode.
FIG. 3 is a frequency response graph for computer simulation results for the slow-DCDC mode that uses the relatively large inductance value of about 1 μH and the relatively large capacitance value of 2.2 μF. The frequency response graph confirms that there are conflicting needs to have a relatively large inductance value and a relatively large decoupling capacitance value for the slow DCDC mode, and a relatively small inductance value and a relatively small capacitance value for the envelope tracking mode. As a result of these conflicting needs, the related art power management system 10 (FIG. 1) cannot effectively supply power to the multi-mode/multi-band PA 12 (FIG. 1) and the bolt-on PAs 14 (FIG. 1) that includes PAs that operate in a saturated mode and that includes PAs that operate in a linear mode.
One related art attempt at solving this conflict is to have a pair of switched inductor and capacitor filter branches that have inductances and capacitances matched to the two different modes of operation. However, the related art switched inductor and capacitor filter branches include series electronic switches that have on state resistance values that lower the quality Q of any filtering provided by the switched inductor and capacitor filter branches. Thus the filtering achieved is not effective enough to pass spectrum requirements. Therefore, a need remains for a power management system that supplies power to a multi-mode/multi-band RF PA load that includes PAs that operate in a saturated mode and that includes PAs that operate in a linear mode.